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1 ? fn6285.0 caution: these devices are sensitive to electrosta tic discharge; follow proper ic handling procedures. 1-888-intersil or 1-888-468-3774 | intersil (and design) is a registered trademark of intersil americas inc. copyright intersil americas inc. 2006. all rights reserved. dynamic vid? is a trademark of intersil americas inc. all other trademarks mentioned are the property of their respective owners. isl6564a multiphase pwm contro ller with linear 6-bit dac capable of precision r ds(on) or dcr differential current sensing the isl6564a is a multiphase pwm controller which controls microprocessor core voltage regulation by driving up to 4 synchronous-rectified buck channels. it features a high bandwidth control loop to provide optimal response to the load transients. with switching freq uency up to 1.5mhz per phase, the isl6564a based voltage regulator requires minimum components and pcb area in dc/dc converter application. the isl6564a senses current by utilizing patented techniques to measure the voltage across the on resistance, r ds(on) , of the lower mosfets or dcr of the output inductor during their conduction intervals. current sensing provides the needed signals for precision droop, channel- current balancing, and overcurrent protection. a unity gain, differential amplifier is provided for remote voltage sensing. any potential difference between remote and local grounds can be completely eliminated using the remote-sense amplifier. eliminating ground differences improves regulation and protec tion accuracy. the threshold- sensitive enable input is available to accurately coordinate the start up of the isl6564a with any other voltage rail. dynamic-vid? technology allows seamless on-the-fly vid changes. the offset pin allows accurate voltage offset settings that are independent of vid setting. the isl6564a uses a 5v bias and has a built-in shunt regulator to allow 12v bias using only a small external limiting resistor. features ? precision multiphase core voltage regulation - differential remote voltage sensing - 0.5% system accuracy - adjustable reference-voltage offset ? precision r ds(on) or dcr current sensing - accurate load-line programming - accurate channel-current balancing - differential current sense - low-cost, lossless current sensing ? internal shunt regulator for 5v or 12v biasing ? microprocessor voltag e identification input - self clocked dynamic vi d? control technology - 6-bit vid input - 0.525v to 1.300v in 12.5mv steps ? threshold-sensitive enable function for power sequencing control ? overcurrent protection ? overvoltage protection - no additional external components needed - ovp pin to drive crowbar device ? 1, 2, 3, or 4 phase operation ? up to 1.5mhz per phase operation (>6mhz ripple) ? qfn package - compliant to jedec pub95 mo-220 qfn - quad flat no leads - product outline - qfn near chip scale package footprint; improves pcb efficiency, thinner in profile ? pb-free plus anneal available (rohs compliant) ordering information part number part marking temp. (c) package pkg. dwg. # isl6564acr isl6564acr 0 to 70 40 ld 6x6 qfn l40.6x6 isl6564acrz (note) isl6564acrz 0 to70 40 ld 6x6 qfn (pb-free) l40.6x6 isL6564AIR isL6564AIR -40 to 85 40 ld 6x6 qfn l40.6x6 isL6564AIRz (note) isL6564AIRz -40 to 85 40 ld 6x6 qfn (pb-free) l40.6x6 add ?-t? suffix for tape and reel. note: intersil pb-free plus anneal products employ special pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish , which are rohs compliant and compatible with both snpb and pb-free soldering operations. intersil pb-free products are msl classi fied at pb-free peak reflow temperatures that meet or exceed the pb-free requirements of ipc/jedec j std-020. data sheet june 21, 2006
2 fn6285.0 june 21, 2006 pinout isl6564a (40 ld qfn) top view vid4 vid3 vid2 vid1 vid0 ofs dac gnd pgood ovp gnd fs drven enll en iout vid5 ref vcc pwm4 isen4+ isen4- isen2- isen2+ pwm2 pwm1 isen1+ isen1- isen3- isen3+ vsen rgnd vdiff comp gnd fb idroop gnd pwm3 1 40 2 3 4 5 6 7 8 9 10 30 29 28 27 26 25 24 23 22 21 39 38 37 36 35 34 33 32 31 11 12 13 14 15 16 17 18 19 20 gnd gnd isl6564a
3 fn6285.0 june 21, 2006 isl6564acr block diagram i_trip ovp latch channel power-on reset (por) pwm1 pwm2 pwm3 pwm4 gnd vcc fb fs s clock and vid5 vid4 vid3 vid2 comp vsen generator sawtooth vid1 rgnd vdiff pgood ovp en 1.29v i_tot dynamic vid d/a sample hold & current balance channel detect ofs three-state ovp vid0 soft-start and fault logic offset ref +200mv r x1 e/a oc pwm pwm pwm pwm q iout enll dac idroop drven isen3- isen4+ isen1+ isen2- channel current sense isen4- isen3+ isen2+ isen1- isl6564a
4 fn6285.0 june 21, 2006 typical application for voltage regulation without droop using r ds(on) sensing vid4 +5v pwm vcc boot ugate phase lgate gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin vid5 pgood vid3 vid2 vid1 vsen vdiff fb comp vcc gnd rgnd en isen1+ pwm1 pwm2 isen2+ pwm3 isen3+ pwm4 isen4+ isl6564a p load vid0 isen1- isen2- isen3- isen4- iout ref dac ovp fs ofs driver isl6612 pvcc enll vidpgood idroop ntc network voltage propotional to load current drven driver isl6612 isl6564a
5 fn6285.0 june 21, 2006 typical application for voltage regul ation without droop using dcr sensing vid4 +5v pwm vcc boot ugate phase lgate gnd +12v driver vin pwm vcc boot ugate phase lgate pvcc gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin vid5 pgood vid3 vid2 vid1 vsen vdiff fb comp vcc gnd rgnd en isen1+ pwm1 pwm2 isen2+ pwm3 isen3+ pwm4 isen4+ isl6564a p load vid0 isen1- isen2- isen3- isen4- iout ref dac ovp fs ofs driver isl6612 pvcc enll vidpgood idroop isl6612 ntc network voltage propotional to load current drven isl6564a
6 fn6285.0 june 21, 2006 typical application for loa d line regulation using r ds(on) sensing and external ntc vid3 +5v pwm vcc boot ugate phase lgate gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin vid4 pgood vid2 vid1 vid0 vsen vdiff fb comp vcc gnd rgnd en isen1+ pwm1 pwm2 isen2+ pwm3 isen3+ pwm4 isen4+ isl6564a p load vid12.5 isen1- isen2- isen3- isen4- iout ref dac ovp fs ofs isl6612 pvcc enll vidpgood idroop isl6612 driver driver +12v ntc network voltage propotional to load current drven ntc thermistor isl6564a
7 fn6285.0 june 21, 2006 typical application for load line regulat ion using dcr sensing and external ntc vid3 +5v pwm vcc boot ugate phase lgate gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin pwm vcc boot ugate phase lgate pvcc gnd +12v isl6612 driver vin vid4 pgood vid2 vid1 vid0 vsen vdiff fb comp vcc gnd rgnd en isen1+ pwm1 pwm2 isen2+ pwm3 isen3+ pwm4 isen4+ isl6564a p load vid12.5 isen1- isen2- isen3- isen4- iout ref dac ovp ntc thermistor fs ofs driver isl6612 pvcc enll vidpgood idroop isl6612 driver +12v ntc network voltage propotional to load current drven isl6564a
8 fn6285.0 june 21, 2006 absolute m aximum ratings supply voltage, vcc . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.5v input and output voltage (except ovp). . gnd -0.3v to v cc + 0.3v ovp voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15v esd (human body model) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>4kv esd (machine model) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>300v esd (charged device model) . . . . . . . . . . . . . . . . . . . . . . . . . .>2kv operating conditions supply voltage, vcc (5v bias mode, note 3) . . . . . . . . . . +5v 5% ambient temperature (isl6564acr, isl6564acrz) . . 0c to 70c ambient temperature (isL6564AIR, isL6564AIRz) . .-40c to 85c thermal information thermal resistance ja (c/w) jc (c/w) qfn package (notes 1, 2). . . . . . . . . . 32 3.5 maximum junction temperature . . . . . . . . . . . . . . . . . . . . . . . 150c maximum storage temperature range . . . . . . . . . . - 65c to 150c maximum lead temperature (soldering 10s) . . . . . . . . . . . . . 300c caution: stress above those listed in ?absolute maximum ratings? may cause permanent damage to the device. this is a stress onl y rating and operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. notes: 1. ja is measured in free air with the component mounted on a high ef fective thermal conductivity te st board with ?direct attach? fe atures. see tech brief tb379. 2. for jc , the ?case temp? location is the center of the exposed metal pad on the package underside. electrical specifications operating conditions: vcc = 5v or icc < 25ma (note 3). unless otherwise specified. parameter test conditions min typ max units vcc supply current nominal supply vcc = 5vdc; en = 5vdc; r t = 100k , isen1 = isen2 = isen3 = isen4 = -70 a -1418ma shutdown supply vcc = 5vdc; en = 0vdc; r t = 100k -1014ma power-on reset and enable por threshold vcc rising 4.20 4.31 4.50 v vcc falling 3.60 3.80 4.00 v enable threshold en rising 1.26 1.29 1.32 v hysteresis 110 125 135 mv fault reset 1.12 1.16 1.20 v enll input logic low level --0.4v enll input logic high level 0.8 - - v enll leakage current enll = 5v - - 1 a reference voltage and dac system accuracy (vid = 1.v-1.3v) (note 4) -0.5 - 0.5 %vid system accuracy (vid = 0.525v-0.9875v) (note 4) -0.9 - 0.9 %vid vid pull-up -55 -45 -35 a vid input low level --0.4v vid input high level 0.8 - - v dac source/sink current vid = 010100 -200 - 200 a vid input voltage when floated 1.0 1.15 1.30 v ref source/sink current -50 - 50 a pin-adjustable offset voltage at ofs pin offset resistor connected to ground 388 400 412 mv vcc = 5.000v, offset resistor connected to vcc 2.91 3.0 3.09 v maximum ofs source and sink current --50 a isl6564a
9 fn6285.0 june 21, 2006 oscillator accuracy r t = 100k 260 300 345 khz adjustment range 0.08 - 1.5 mhz sawtooth amplitude -2- v max duty cycle - 66.7 - % error amplifier open-loop gain r l = 10k to ground - 80 - db open-loop bandwidth c l = 100pf, r l = 10k to ground - 18 - mhz maximum output voltage 4.0 4.3 - v output high voltage @ 2ma 3.7 - - v output low voltage @ 2ma - - 1.35 v remote-sense amplifier bandwidth -20-mhz output voltage @ 1ma load vsen - rgnd = 2.5v 2.48 2.50 2.52 v pwm output pwm output voltage low iload = 500 a--0.5v pwm output voltage high iload = 500 a4.0--v driver enable output drven output voltage low iload = 1ma - - 0.3 v drven output voltage high iload = 1ma 4.0 - - v sense current output sensed current accuracy isen1 = isen2 = isen3 = isen4 = 80 a 758593 a overcurrent trip level 95 110 125 a maximum voltage at idroop and iout vcc = 4.5v (note 5) 2 v power good and protection monitors pgood low voltage i pgood = 4ma - - 0.3 v undervoltage offset from vid vsen falling 70 75 80 %vid overvoltage threshold voltage above vid, after soft-start (note 6) 180 200 230 mv before enable 1.40 1.45 1.50 v vcc < por threshold 1.7 1.8 1.87 v overvoltage reset voltage vcc por threshold, vsen falling - 0.6 - v vcc < por threshold - 1.5 - v ovp drive voltage i ovp = -10ma, vcc = 5v 3.0 3.6 5.0 v notes: 3. when using the internal shunt regulat or, vcc is clamped to 6.2v (max). cu rrent must be limited to 25ma or less. 4. these parts are designed and adjusted for accuracy with all errors in the voltage loop included. 5. guaranteed by design. 6. during soft-start, vdac rises from 0 to vid. the over voltage trip level is the higher of 1.5v and vdac + 0.2v. electrical specifications operating conditions: vcc = 5v or icc < 25ma (note 3). unless otherwise specified. (continued) parameter test conditions min typ max units isl6564a
10 fn6285.0 june 21, 2006 functional pin description vcc supplies all the power necessary to operate the chip. the controller starts to operate when the voltage on this pin exceeds the rising por threshold and shuts down when the voltage on this pin drops below the falling por threshold. connect this pin directly to a +5v supply or through a series 300 resistor to a +12v supply. gnd bias and reference ground for the ic. en this pin is a threshold-se nsitive enable input for the controller. connecting the 12v supply to en through an appropriate resistor divider provides a means to synchronize power-up of the controller and the mosfet driver ics. when en is driven above 1.29v, the isl6564a is active depending on status of enll, the internal por, and pending fault states. driving en below 1.16v will clear all fault states and prime the isl6564a to soft-start when re-enabled. enll this pin is a logic-level enable input for the controller. when asserted to a logic high, the isl6564a is active depending on status of en, the internal por, vid inputs and pending fault states. deasserting enll will clear all fault states and prime the isl6564a to soft-start when re-enabled. when floating, enll pin will be pulled to high internally with a typical voltage as 1.15v. fs a resistor, rt, placed from fs to ground will set the switching frequency. there is an inverse relationship between the value of the resistor and the switching frequency. see figure 20 and equation 30. vid5, vid4, vid3, vid2, vid1, and vid0 these are the inputs to the in ternal dac that provides the reference voltage for output regulation. connect these pins either to open-drain outputs with or without external pull-up resistors or to active-pull-up outputs. vid5-vid0 have 45a internal pull-up current sources that diminish to zero as the voltage rises above the logic-high level. these inputs can be pulled up as high as vcc plus 0.3v. vdiff, vsen, and rgnd vsen and rgnd form the precis ion differential remote- sense amplifier. this amplifier converts the differential voltage of the remote output to a single-ended voltage referenced to local ground. vdiff is the amplifier?s output and the input to the regulation and protection circuitry. connect vsen and rgnd to the sense pins of the remote load. fb and comp inverting input and output of the error amplifier, respectively. fb is connected to vdiff through a resistor. a negative current, proportional to output current is present on the fb pin. a properly sized resistor between vdiff and fb sets the load line (droop). the droop scale factor is set by the ratio of the isen resistors and the lower mosfet r ds(on) . comp is tied back to fb through an external r-c network with no dc connection to compensate the regulator. dac and ref the dac output pin is the outpu t of the precision internal dac reference. the ref input pin is the positive input of the error amp. in typical applications, a 1k , 1% resistor is used between dac and ref to generat e a precise offset voltage. this voltage is proportional to the offset current determined by the offset resistor from ofs to ground or vcc. a capacitor is used between ref and ground to smooth the voltage transition during dy namic vid? operations. pwm1, pwm2, pwm3, pwm4 pulse-width modulation outputs. connect these pins to the pwm input pins of the inters il driver ic. the number of active channels is determined by the state of pwm3 and pwm4. leave pwm4 unconnected and tie pwm3 to vcc to configure for 2-phase operation. tie pwm4 to vcc to configure for 3-phase operation. tie both pwm4 and pwm3 to high for 1-phase operation. isen1+, isen1-; isen2+, isen2-; isen3+, isen3-; isen4+, isen4- the isen+ and isen- pins are current sense inputs to individual differential amplifiers. the sensed current is used as a reference for channel balancing, protection, and regulation. inactive channels should have their respective sense inputs left open (for example, for 3-phase operation open isen4+). for dcr sensing, connect each isen- pin to the node between the rc sense elements. tie the isen+ pin to the other end of the sense capacitor through a resistor, r isen . the voltage across the sense capacitor is proportional to the inductor current. the sensed current is proportional to the output current, and scaled by the dcr of the inductor, divided by r isen . when configured for r ds(on) current sensing, the isen1-, isen2-, isen3-, and isen4- pins are grounded at the lower mosfet sources. the isen 1+, isen2+, isen3+, and isen4+ pins are then held at a virtual ground, such that a resistor connected between them, and the drain terminal of the associated lower mosf et, will carry a current proportional to the current flowing through that channel. the current is determined by the negative voltage developed across the lower mosfet?s r ds(on) , which is the channel current scaled by r ds(on) . isl6564a
11 fn6285.0 june 21, 2006 pgood pgood is used as an indication of the end of soft-start per the microprocessor specification. it is an open-drain logic output that is low impedance until the soft-start is completed. it will be pulled low again once the undervoltage point is reached. ofs the ofs pin provides a means to program a dc offset current for generating a dc offs et voltage at the ref input. the offset current is generated via an external resistor and precision internal voltage references. the polarity of the offset is selected by connecting the resistor to gnd or vcc. for no offset, the ofs pin should be left unterminated. ovp overvoltage protection pin. this pin pulls to vcc and is latched when an overvoltage condition is detected. connect this pin to the gate of an scr or mosfet tied from v in or v out to ground to prevent damage to the load. this pin may be pulled above vcc as high as 15v to ground with an external resistor. however, it is only capable of pulling low when vcc is above 2v. drven driver enable pin. this pin can be used to enable the drivers which have enable pins such as isl6605 or isl6608. if isl6564a is used with intersil isl6612 drivers, it?s not necessary to use this pin. idroop and iout idroop and iout are the out put pins of sensed average channel current which is proportional to load current. they are designed for flexible application purposes. in the application which does not require loadline, leave idroop pin open. in the applic ation which requires load line, connect idroop pin to fb so that the sensed average current will flow through the resistor between fb and vdiff to create a voltage drop which is proportional to load current. iout is typically used for load current indication. operation multiphase power conversion microprocessor load current pr ofiles have changed to the point that the advantages of multiphase power conversion are impossible to ignore. the technical challenges associated with producing a single-phase converter which is both cost-effective and thermally viable have forced a change to the cost-saving approach of multiphase. the isl6564a controller helps reduce the complexity of implementation by integrating vital functions and requiring minimal output components. the block diagrams on pages 3, 4, 5, 6, and 7 provide top level views of multiphase power conversion using the isl6564a controller. interleaving the switching of each channel in a multiphase converter is timed to be symmetrically out of phase with each of the other channels. in a 3-phase converter, each channel switches 1/3 cycle after the previous channel and 1/3 cycle before the following channel. as a result, the three-phase converter has a combined ripple frequency three times greater than the ripple frequency of any one phase. in addition, the peak-to- peak amplitude of the combined inductor currents is reduced in proportion to the number of phases (equations 1 and 2). increased ripple frequency and lower ripple amplitude mean that the designer can use less per-channel inductance and lower total output capacitance for any performance specification. figure 1 illustrates the multiplic ative effect on output ripple frequency. the three channel currents (il1, il2, and il3) combine to form the ac ripple current and the dc load current. the ripple component has three times the ripple frequency of each individual channel current. each pwm pulse is terminated 1/3 of a cycle after the pwm pulse of the previous phase. the peak-to- peak current for each phase is about 7a, and the dc components of the inductor currents combine to feed the load. to understand the reduction of ripple current amplitude in the multiphase circuit, examine the equation representing an individual channel?s peak-to-peak inductor current. in equation 1, v in and v out are the input and output voltages respectively, l is the single-channel inductor value, and f s is the switching frequency. figure 1. pwm and inductor-current waveforms for 3-phase converter 1s/div pwm2, 5v/div pwm1, 5v/div il2, 7a/div il1, 7a/div il1 + il2 + il3, 7a/div il3, 7a/div pwm3, 5v/div i pp v in v out ? () v out lf s v in ----------------------------------------------------- - = (eq. 1) isl6564a
12 fn6285.0 june 21, 2006 the output capacitors conduct the ripple component of the inductor current. in the case of multiphase converters, the capacitor current is the sum of the ripple currents from each of the individual channels. compare equation 1 to the expression for the peak-to-peak current after the summation of n symmetrically phase-shi fted inductor currents in equation 2. peak-to-peak ripple current decreases by an amount proportional to the number of channels. output- voltage ripple is a function of capacitance, capacitor equivalent series resistance (esr), and inductor ripple current. reducing the inductor ripple current allows the designer to use fewer or less costly output capacitors. another benefit of interleaving is to reduce input ripple current. input capacitance is determined in part by the maximum input ripple current. multiphase topologies can improve overall system cost and size by lo wering input ripple current and allowing the designer to reduce the cost of input capacitance. the example in figure 2 illustrates input currents from a three-phase converter combining to reduce the total input ripple current. the converter depicted in figure 2 delivers 36a to a 1.5v load from a 12v input. the rms inpu t capacitor current is 5.9a. compare this to a single-phase converter also stepping down 12v to 1.5v at 36a. the sing le-phase converter has 11.9a rms input capacitor current. the single-phase converter must use an input capacitor bank with twice the rms current capacity as the equivalent three-phase converter. figures 21, 22 and 23 in the section entitled input capacitor selection can be used to determine the input-capacitor rms current based on load current, duty cycle, and th e number of channels. they are provided as aids in determining the optimal input capacitor solution. figure 24 shows the single phase input-capacitor rms current for comparison. pwm operation the timing of each converter leg is set by the number of active channels. the default channel setting for the isl6564a is four. one switching cycle is defined as the time between pwm1 pulse termination signals. the pulse termination signal is an internally generated clock signal which triggers the falling edge of pwm1. the cycle time of the pulse termination signal is the inverse of the switching frequency set by the resistor between the fs pin and ground. each cycle begins when the clock signal commands the channel 1 pwm output to go low. the pwm1 transition signals the channel-1 mosfet driver to turn off the channel 1 upper mosfet and turn on the channel 1 synchronous mosfet. in the default channel configuration, the pwm2 pulse terminates 1/4 of a cycle after pwm1. the pwm3 output follows another 1/4 of a cycle after pwm2. pwm4 terminates another 1/ 4 of a cycle after pwm3. if pwm3 is connected to vcc, two channel operation is selected and the pwm2 pulse te rminates 1/2 of a cycle later. connecting pwm4 to vcc selects three channel operation and the pulse-termination ti mes are spaced in 1/3 cycle increments. connecting both pwm3 and pwm4 to vcc selects single-channel operation. once a pwm signal transitions low, it is held low for a minimum of 1/3 cycle. this forc ed off time is required to ensure an accurate current sample. current sensing is described in the next section. after the forced off time expires, the pwm output is enab led. the pwm output state is driven by the position of th e error amplifier output signal, v comp , minus the current correction signal relative to the sawtooth ramp as illustrated in figure 7. when the modified v comp voltage crosses the sawtoot h ramp, the pwm output transitions high. the mosfet driver detects the change in state of the pwm signal and turns off the synchronous mosfet and turns on the upper mosfet. the pwm signal will remain high until the pulse termination signal marks the beginning of the next cycle by triggering the pwm signal low. current sampling during the forced off-time fo llowing a pwm transition low, the associated channel current sense amplifier uses the isen inputs to reproduce a signal proportional to the inductor current, i l . no matter the current sense method, the sense current, i sen , is simply a scaled version of the inductor current. coincident with the falling edge of the pwm signal, the sample and ho ld circuitry samples i sen , as illustrated in figure 3. the sample window hold time, t hold , is fixed and equal to 1/3 of the switching period, t sw . therefore, the sample current, i n , is proportional to the output current and held for one switching cycle. the sample current is used for current balance, load-line regulation, and overcurrent protection. figure 2. channel input currents and input- capacitor rms current for 3-phase converter channel 1 input current 10a/div channel 2 input current 10a/div channel 3 input current 10a/div input-capacitor current, 10a/div 1s/div i cpp , v in nv out ? () v out lf s v in ----------------------------------------------------------- - = (eq. 2) t hold t sw 3 ---------- 1 3f sw ? ------------------ == (eq. 3) isl6564a
13 fn6285.0 june 21, 2006 current sensing the isl6564a supports inductor dcr sensing, mosfet r ds(on) sensing, or resistive sensing techniques. the internal circuitry, shown in figures 4, 5, and 6, represents channel n of an n-channel converter. this circuitry is repeated for each channel in the converter, but may not be active depending on the status of the pwm3 and pwm4 pins, as described in the pwm operation section. inductor dcr sensing an inductor?s winding is characteristic of a distributed resistance as measured by the dcr (direct current resistance) parameter. consider the inductor dcr as a separate lumped quantity, as shown in figure 4. the channel current i l , flowing through the inductor, will also pass through the dcr. equation 4 shows the s-domain equivalent voltage across the inductor v l . a simple r-c network across the inductor extracts the dcr voltage, as shown in figure 4. the voltage on the capacitor v c , can be shown to be proportional to the channel current i l , see equation 5. if the r-c network components are selected such that the rc time constant matches the inductor l/dcr time constant, then v c is equal to the voltage drop across the dcr. the capacitor voltage v c , is then replicated across the sense resistor r isen . the current through the sense resistor is proportional to the inductor current. equation 6 shows the proportion between the chann el current and the sensed current i sen , is driven by the valu e of the sense resistor chosen and the dcr of the inductor. dcr varies with temperature, so a positive temperature coefficient (ptc) resistor should be selected for the sense resistor r isen . resistive sensing if dcr sensing is not utilized, independent current-sense resistors in series with each out put inductor can serve as the sense element (see figure 5). this technique is more accurate, but reduces overall converter efficiency due to the addition of a lossy element directly in the output path. figure 3. sample and hold timing time pwm i l switching period i sen t hold sample current, i n v l i l sl dcr + ? () ? = (eq. 4) v c s l dcr ------------- ? 1 + ?? ?? dcr i l ? () ? src 1 + ? () -------------------------------------------------------------------- - = (eq. 5) figure 4. dcr sensing configuration i n i sen i l dcr r isen ----------------- - = - + isen-(n) sample & hold isl6564a internal circuit v in isen+(n) pwm(n) isl6207 r isen(n) dcr l inductor r v out c out (ptc) - + v c (s) c i l s () - + v l i sen i l dcr r isen ----------------- - ? = (eq. 6) isl6564a
14 fn6285.0 june 21, 2006 mosfet r ds(on) sensing the controller can also sense the channel load current by sampling the voltage across the lower mosfet r ds(on) (see figure 6). the amplifier is ground-reference by connecting the isen- input to the source of the lower mosfet. isen+ connects to the phase node through a resistor r isen . the voltage across r isen is equivalent to the voltage drop across the r ds(on) of the lower mosfet while it is conducting. the re sulting current into the isen+ pin is proportional to the channel current i l . the isen current is then sampled and held after sufficient settling time. the sampled current i n , is used for channel-current balance, load-line regulation, and overcurrent protection. from figure 6, equation 7 for i sen is derived. where i l is the channel current. since mosfet r ds(on) increases with temperature, a ptc resistor should be chosen for r isen to compensate for this change. channel-current balance the sampled currents i n , from each active channel are summed together and divided by the number of active channels. the resulting cycle average current i avg , provides a measure of the total load current demand on the converter during each switching cycle. channel current balance is achieved by comparing the samp led current of each channel to the cycle average current, and making an appropriate adjustment to each channel pulse width based on the error. intersil?s patented current-balanc e method is illustrated in figure 7, with error correction for channel 1 represented. in the figure, the cycle average current combines with the channel 1 sample, i 1 , to create an error signal i er . the filtered error signal modifies the pulse width commanded by v comp to correct any unbalance and force i er toward zero. the same method for error signal correction is applied to each active channel. channel current balance is esse ntial in realizing the thermal advantage of multiphase operatio n. the heat generated in down converting is dissipated over multiple devices and a greater area. the designer avoids the complexity of driving multiple parallel mosfets, and the expense of using heat sinks and nonstandard magnetic materials. voltage regulation the integrating compensation network shown in figure 8 assures that the steady-state error in the output voltage is limited only to the error in the reference voltage (output of the dac) and offset errors in the ofs current source, remote-sense and error amplifiers. intersil specifies the guaranteed tolerance of the isl6564a to include the combined tolerances of each of these elements. the output of the error amplifier, v comp , is compared to the sawtooth waveform to generate the pwm signals. the pwm signals control the timing of the intersil mosfet drivers and regulate the converter output to the specified reference voltage. the internal and external circuitry which control voltage regulation is illustrated in figure 8. figure 5. sense resistor in series with inductors i n i sen i l r sense r isen -------------------------- = - + isen-(n) sample & hold isl6564a internal circuit isen+(n) r isen(n) r sense l v out c out i l figure 6. mosfet r ds(on) current-sensing circuit i n i sen i l r ds on () r isen ------------------------- - = - + isen+(n) r isen sample & hold isl6564a internal circuit external circuit v in n-channel mosfets - + i l r ds on () i l isen-(n) (ptc) i sen i l r ds on () r isen ---------------------- = (eq. 7) figure 7. channel 1 pwm function and current- balance adjustment n i avg i 4 * i 3 * i 2 - + + - + - f(j ) pwm1 i 1 v comp sawtooth signal i er note: *channels 3 and 4 are optional. filter isl6564a
15 fn6285.0 june 21, 2006 the isl6564a incorporates an internal differential remote- sense amplifier in the feedback path. the amplifier removes the voltage error encountered when measuring the output voltage relative to the local controller ground reference point resulting in a more accurate means of sensing output voltage. connect the microprocessor sense pins to the non-inverting input, vsen, and in verting input, rgnd, of the remote-sense amplifier. the remote-sense output, v diff , is connected to the inverting input of the error amplifier through an external resistor. a digital to analog converter (dac) generates a reference voltage based on the state of logic signals at pins vid4 through vid12.5. the dac decodes the a 6-bit logic signal (vid) into one of the discrete voltages shown in table 1. each vid input offers a 45 a pull-up to an internal 2.5v source for use with open-drain outputs. the pull-up current diminishes to zero above the logic threshold to protect voltage-sensitive output devices. external pull-up resistors can augment the pull-up current sources if case leakage into the driving device is greater than 45 a. load-line regulation some microprocessor manufact urers require a precisely- controlled output resistance. this dependence of output voltage on load current is often termed ?droop? or ?load line? regulation. by adding a well controlled output impedance, the output voltage can effectively be level shifted in a direction which works to achieve the load-line regulation required by these manufacturers. figure 8. output voltage and load-line regulation with offset adjustment i avg external circuit isl6564a internal circuit comp r c r fb fb vdiff vsen rgnd - + v droop error amplifier - + v out + differential remote-sense amplifier v comp c c ref dac r ref c ref - + v out - idroop table 1. voltage identification (vid) codes vid5 vid4 vid3 vid2 vid1 vid0 vdac 400 mv 200 mv 100 mv 50 mv 25 mv 12.5 mv 1111 1 1 off 1 1 1 1 1 0 1.3000v 1 1 1 1 0 1 1.2875v 1 1 1 1 0 0 1.2750v 1 1 1 0 1 1 1.2625v 1 1 1 0 1 0 1.2500v 1 1 1 0 0 1 1.2375v 1 1 1 0 0 0 1.2250v 1 1 0 1 1 1 1.2125v 1 1 0 1 1 0 1.2000v 1 1 0 1 0 1 1.1875v 1 1 0 1 0 0 1.1750v 1 1 0 0 1 1 1.1625v 1 1 0 0 1 0 1.1500v 1 1 0 0 0 1 1.1375v 1 1 0 0 0 0 1.1250v 1 0 1 1 1 1 1.1125v 1 0 1 1 1 0 1.1000v 1 0 1 1 0 1 1.0875v 1 0 1 1 0 0 1.0750v 1 0 1 0 1 1 1.0625v 1 0 1 0 1 0 1.0500v 1 0 1 0 0 1 1.0375v 1 0 1 0 0 0 1.0250v 1 0 0 1 1 1 1.0125v 1 0 0 1 1 0 1.0000v 1 0 0 1 0 1 0.9875v 1 0 0 1 0 0 0.9750v 1 0 0 0 1 1 0.9625v 1 0 0 0 1 0 0.9500v 1 0 0 0 0 1 0.9375v 1 0 0 0 0 0 0.9250v 0 1 1 1 1 1 0.9125v 0 1 1 1 1 0 0.9000v 0 1 1 1 0 1 0.8875v 0 1 1 1 0 0 0.8750v isl6564a
16 fn6285.0 june 21, 2006 in other cases, the designer may determine that a more cost-effective solution can be achieved by adding droop. droop can help to reduce the output-voltage spike that results from fast load-current demand changes. the magnitude of the spike is dictated by the esr and esl of the output capacitors selected. by positioning the no-load voltage level near the upper s pecification limit, a larger negative spike can be sustained without crossing the lower limit. by adding a well cont rolled output impedance, the output voltage under load can effectively be level shifted down so that a larger positive spike can be sustained without crossing the upper specification limit. as shown in figure 8, a current proportional to the average current in all active channels, i avg , flows from fb through a load-line regulation resistor, r fb . the resulting voltage drop across r fb is proportional to the output current, effectively creating an output voltage dro op with a steady-state value defined as the regulated output voltage is reduced by the droop voltage v droop . the output voltage as a function of load current is derived by combining equation 8 with the appropriate sample current expression defined by the current sense method employed. where v ref is the reference voltage, v ofs is the programmed offset voltage, i out is the total output current of the converter, r isen is the sense resistor in the isen line, and r fb is the feedback resistor. r x has a value of dcr, r ds(on) , or r sense depending on the sensing method. output-voltage offset programming the isl6564a allows the designer to accurately adjust the offset voltage. when a resistor, r ofs , is connected between ofs to vcc, the voltage across it is regulated to 2.0v. this causes a proportional current (i ofs ) to flow into ofs. if r ofs is connected to ground, the voltage across it is regulated to 0.5v, and i ofs flows out of ofs. a resistor between dac and ref, r ref , is selected so that the product (i ofs x r ofs ) is equal to the desired offset voltage. these functions are shown in figure 9. as it may be noticed in figure 9, the ofsout pin must be connected to the ref pin for this current injection to function in isl6564acr. the current flow through r ref creates an offset at the ref pin, which is ultimately duplicated at the output of the regulator. once the desired output offset voltage has been determined, use the following formulas to set r ofs : for positive offset (connect r ofs to vcc): for negative offset (connect r ofs to gnd): 0 1 1 0 1 1 0.8625v 0 1 1 0 1 0 0.8500v 0 1 1 0 0 1 0.8375v 0 1 1 0 0 0 0.8250v 0 1 0 1 1 1 0.8125v 0 1 0 1 1 0 0.8000v 0 1 0 1 0 1 0.7875v 0 1 0 1 0 0 0.7750v 0 1 0 0 1 1 0.7625v 0 1 0 0 1 0 0.7500v 0 1 0 0 0 1 0.7375v 0 1 0 0 0 0 0.7250v 0 0 1 1 1 1 0.7125v 0 0 1 1 1 0 0.7000v 0 0 1 1 0 1 0.6875v 0 0 1 1 0 0 0.6750v 0 0 1 0 1 1 0.6625v 0 0 1 0 1 0 0.6500v 0 0 1 0 0 1 0.6375v 0 0 1 0 0 0 0.6250v 0 0 0 1 1 1 0.6125v 0 0 0 1 1 0 0.6000v 0 0 0 1 0 1 0.5875v 0 0 0 1 0 0 0.5750v 0 0 0 0 1 1 0.5625v 0 0 0 0 1 0 0.5500v 0 0 0 0 0 1 0.5375v 0 0 0 0 0 0 0.525v table 1. voltage identification (vid) codes (continued) vid5 vid4 vid3 vid2 vid1 vid0 vdac v droop i avg r fb = (eq. 8) v out v ref v offset ? i out 4 ------------- r x r isen ----------------- -r fb ?? ?? ?? ? = (eq. 9) r ofs 2r ref v offset -------------------------- = (eq. 10) r ofs 0.5 r ref v offset ----------------------------- - = (eq. 11) isl6564a
17 fn6285.0 june 21, 2006 dynamic vid modern microprocessors need to make changes to their core voltage as part of normal operation. they direct the core-voltage regulator to do this by making changes to the vid inputs during regulator operation. the power management solution is required to monitor the dac inputs and respond to on-the-fly vid changes in a controlled manner. supervising the safe ou tput voltage transition within the dac range of the proce ssor without discontinuity or disruption is a necessary function of the core-voltage regulator. the isl6564a checks the vid inputs at the three edges of 16mhz clock. if the vid code is found to have changed, the controller waits half of a co mplete cycle befo re executing a 12.5mv change. if during the ha lf-cycle wait period, the difference between dac level and the new vid code changes sign, no change is made. if the vid code is more than 1 bit higher or lower than the dac (not recommended), the controller will execute step-up and step down vid change at a speed of 12.5mv every 4 s until vid and dac are equal. in order to ensure the smooth transition of output voltage during vid change, a vid step change smoothing network composed of r ref and c ref is required for an isl6564a based voltage regulator. the selection of r ref is based on the desired offset as detailed above in output-voltage offset programming . the selection of c ref is based on the time duration for 1 bit vid change and the allowable delay time. assuming the microprocessor co ntrols the vid change at 1 bit every t vid , the relationship between the time constant of r ref and c ref network and t vid is given by equation 12. where, t vid = 4 s, k is the number of the internal vid change cycle. if typically r ref is selected to be 1k , the allowable delay time for vr to respond to new vid code is 5 vid change cycles (totally 20 s), the value of c ref should be 22nf based on equation 12. operation initialization prior to converter initialization, proper conditions must exist on the enable inputs and vcc. when the conditions are met, the controller begins soft-sta rt. once the output voltage is within the proper window of operation, pgood asserts logic 1. enable and disable while in shutdown mode, the pwm outputs are held in a high-impedance state to assure the drivers remain off. the following input conditions must be met before the isl6564a is released from shutdown mode. 1. the bias voltage applied at vcc must reach the internal power-on reset (por) rising threshold. once this threshold is reached, proper operation of all aspects of the isl6564a is guaranteed . hysteresis between the rising and falling thresholds assure that once enabled, the isl6564a will not inadvertently turn off unless the bias voltage drops substantially (see electrical specifications ). dynamic vid d/a e/a vcc dac fb ref ofs vcc gnd + - + - 0.5v 2.0v or gnd r ofs r ref isl6564acr figure 9. output voltag e offset programming with isl6564acr c ref r ref kt vid = (eq. 12) figure 10. power sequencing using threshold- sensitive enable (en) function - + 1.23v external circuit isl6564a internal circuit en +12v por circuit 10.7k 1.40k enable comparator soft-start and fault logic enll vcc isl6564a
18 fn6285.0 june 21, 2006 2. the isl6564a features an enable input (en) for power sequencing between the controller bias voltage and another voltage rail. the enable comparator holds the isl6564a in shutdown until the voltage at en rises above 1.29v. the enable comparat or has about 125mv of hysteresis to prevent bounce . it is important that the driver ics reach their por level before the isl6564a becomes enabled. the schematic in figure 10 demonstrates sequencing the isl6564a with the isl66xx family of intersil mo sfet drivers, which require 12v bias. 3. the voltage on enll must be logic high to enable the controller. this pin is ty pically connected to the vid_pgood. 4. the vid code must not be 111111. this code signals the controller that no load is present. the controller will enter shut-down mode after receiving this code and will execute soft-start upon rece iving any other code. this code can be used to enable or disable the controller but it is not recommended. after receiving this code, the controller executes a 2-cycl e delay before changing the overvoltage trip level to the shut-down level and disabling pwm. overvoltage shutdown can not be reset using this code. to enable the controller, vcc must be greater than the por threshold; the voltage on en must be greater than 1.29v; for isl6564acr, enll must be logic high; and vid cannot be equal to 111111. w hen each of these conditions is true, the controller immediately begins the soft-start sequence. soft-start during soft-start, the dac volt age ramps linearly from zero to the programmed vid level as shown in figure 11. the pwm signals remain in the high-impedance state until the controller detects that the ramping dac level has reached the pre-bias output-v oltage level. this protects the system against the large, negative inductor currents that would otherwise occur when starting with a pre-existing charge on the output as the controller attempted to regulate to zero volts at the beginning of the so ft-start cycle. the soft-start time, t ss , begins with a delay period equal to 64 switching cycles followed by a linear ramp with a fixed rate at a speed of 12.5mv/32s. equation 13 can be used to calculate the soft-start time. for example, when vid is set to 1.2v, the soft-start time will be 3.072ms. a 100mv offset exists on the remote-sense amplifier at the beginning of soft-start and ramps to zero during the first 640 cycles of soft-start (704 cycles following enable). this prevents the large inrush current that would otherwise occur should the output voltage star t out with a slight negative bias. fault monitoring and protection the isl6564a actively monitors output voltage and current to detect fault conditions. fault monitors trigger protective measures to prevent damage to a microprocessor load. one common power good indicator is provided for linking to external system monitors. the schematic in figure 12 outlines the interaction between the fault monitors and the power good signal. power good signal the power good pin (pgood) is an open-drain logic output indication that the converter is operating after soft-start. pgood pulls low during shutdown and releases high after a successful soft-start. pgood wil l only transition low when an undervoltage condition is det ected or the controller is disabled by a reset from en , enll, por, or one of the no-cpu vid codes. after an undervoltage event, pgood will return high unless the controller has been disabled. pgood does not automatically transition low upon detection of an overvoltage condition. t ss 2560 () vid = (eq. 13) figure 11. soft-start waveforms with an un-biased output vout, 500mv/div en, 5v/div 500s/div isl6564a
19 fn6285.0 june 21, 2006 undervoltage detection the undervoltage threshold is set at 75% of the vid code. when the output vo ltage at vsen is bel ow the undervoltage threshold, pgood gets pulled low. overvoltage protection when vcc is above 1.4v, but otherwise not valid as defined under power-on reset in electrical specifications , the overvoltage trip circuit is active using auxiliary circuitry. in this state, an overvo ltage trip occurs if the voltage at vsen exceeds 1.8v. with valid vcc, the overvolta ge circuit is sensitive to the voltage at vdiff. in this state, the trip level is 1.7v prior to valid enable conditions being met as described in enable and disable . the only exception to this is when the ic has been disabled by an overvoltage trip. in that case the overvoltage trip point is vid plus 200mv. during soft-start, the overvoltage trip level is th e higher of 1.5v or vid plus 200mv. upon successful soft-start, the overvoltage trip level is 200mv above vid. two ac tions are taken by the isl6564a to protect the microprocessor load when an overvoltage condition occurs. at the inception of an overvoltage event, all pwm outputs are commanded low instantly (less than 20ns) until the voltage at vsen falls below 0. 6v with valid vcc or 1.5v otherwise. this causes the intersil drivers to turn on the lower mosfets and pull the output voltage below a level that might cause damage to the load. the pwm outputs remain low until vdiff falls to the programmed dac level when they enter a high-impedance state. the intersil drivers respond to the high-impedance input by turning off both upper and lower mosfets. if the overvoltage condition reoccurs, the isl6564a will again command the lower mosfets to turn on. the isl6564a will continue to protect the load in this fashion as long as the overvoltage condition recurs. simultaneous to the protective action of the pwm outputs, the ovp pin pulls to vcc deliv ering up to 100ma to the gate of a crowbar mosfet or scr plac ed either on the input rail or the output rail. turning on the mosfet or scr collapses the power rail and causes a fuse placed further up stream to blow. the fuse must be sized such that the mosfet or scr will not overheat before the fuse blows. the ovp pin is tolerant to 12v (see absolute maximum ratings ), so an external resistor pull-up can be used to augment the driving capability. if using a pull-up resistor in conjunction with the internal overvoltage protection function, care must be taken to avoid nuisance trips that could occur when vcc is below 2v. in that case, the controller is incapable of holding ovp low. once an overvoltage condition is detected, normal pwm operation ceases until the isl6564a is reset. cycling the voltage on en or enll or vcc below the por-falling threshold will reset the controller. cycling the vid codes will not reset the controller. overcurrent protection isl6564a has two levels of ov ercurrent protection. each phase is protected from a sust ained overcurrent condition on a delayed basis, while the combined phase currents are protected on an instantaneous basis. in instantaneous protection mode, the isl6564a takes advantage of the proportionality between the load current and the average current, i avg , to detect an overcurrent condition. see the channel-current balance section for more detail on how the average current is measured. the average current is continually compared with a constant 110 a reference current as shown in figure 12. once the average current exceeds the reference current, a comparator triggers the converter to shutdown. in individual overcurrent pr otection mode, the isl6564a continuously compares the curre nt of each channel with the same 110 a reference current. if any channel current exceeds the reference current continuously for eight consecutive cycles, the comparator triggers the converter to shutdown. figure 12. power good and protection circuitry ovp - + vid + 0.2v vdiff - + 110a i avg - + dac reference ov oc uv pgood 75% soft-start, fault and control logic - + oc i 1 repeat for each channel 110a isl6564a
20 fn6285.0 june 21, 2006 at the beginning of overcurrent shutdown, the controller places all pwm signals in a high-impedance state within 20ns commanding the intersil mosfet driver ics to turn off both upper and lower mosfets. the system remains in this state a period of 4096 switching cycles. if the controller is still enabled at the end of this wait period, it will attempt a soft-start. if the fault remain s, the trip-retry cycles will continue indefinitely (as shown in figure 13) until either controller is disabled or the fa ult is cleared. note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there, there is no thermal hazard during this kind of operation. driver enable output the isl6564a has a driver enable output pin drven. the drven is designed for the application where isl6564a needs to work with drivers that can not recognize three-state pwm input. in normal operation, drven remains low until isl6564a begins soft-start ramp and then changes to high (figure 14). when an overcurrent event occurs, drven is pulled to low instantly (less than 20ns) to disable the driver so that both upper and lower fets be turned off (figure 15). during an overvoltage condition, drven remains high to allow the driver turn on the lower fets based on the pwm input to discharge the energy stored in the output inductor. once the output voltage is reduced to 0.6v, drven is pulled to low as shown in figure 16. there?s no need to use drven when isl6564a is used to work with intersil?s drivers such as isl6612 and isl6605. current sense output the isl6564a has 2 current s ense output pins idroop and iout. they are identical. in ty pical application, idroop pin is connected to fb pin for the application where load line is required. iout pin was designed for load current measurement. as shown in typical application schematics 0a 0v 2ms/div output current, 50a/div figure 13. overcurrent behavior in hiccup mode, f sw = 500khz output voltage, 500mv/div figure 14. drven waveform at start-up vout, 1v/div en, 5v/div 500s/div drven, 5v/div 0a 0v 2ms/div output current, 50a/div figure 15. drven during overcurrent operation output voltage, 500mv/div drven, 5v/div 0v 2ms/div figure 16. drven during overcurrent operation output voltage, 500mv/div drven, 5v/div isl6564a
21 fn6285.0 june 21, 2006 on pages 4 to 7, load current information can be obtained by measuring the voltage between iout to ground when a ntc network from iout pin to the ground is placed. the output current at iout pin is propor tional to load current as shown in figure 17. when selecting the equivalent resistor network components values, it is important to ensure the voltage at iout pin not exceed 2v. when isl6564a is operated at single phase mode (both pwm3 and pwm4 connected to vcc and pwm2 unconnected). the output current at iout and idroop is half of the sensed phase current. general design guide this design guide is intended to provide a high-level explanation of the steps neces sary to create a multiphase power converter. it is assumed that the reader is familiar with many of the basic skills and te chniques referenced below. in addition to this guide, intersil provides complete reference designs that include schematics, bills of materials, and example board layouts for all common microprocessor applications. power stages the first step in designing a multiphase converter is to determine the number of phases. this determination depends heavily on the cost analysis which in turn depends on system constraints that differ from one design to the next. principally, the designer will be concerned with whether components can be mounted on both sides of the circuit board; whether through-hole components are permitted; and the total board space available for power-supply circuitry. generally speaking, the most economical solutions are those in which each phase handles between 15 and 20a. all surface-mount designs will tend toward the lower end of this current range. if through-hole mosfets and inductors can be used, higher per-phase currents are possible. in cases where board space is the limiting constraint, current can be pushed as high as 40a per phase, but these designs require heat sinks and forced air to cool the mosfets, inductors and heat-dissipating surfaces. mosfets the choice of mosfets depends on the current each mosfet will be required to conduct; the switching frequency; the capability of the mosfets to dissipate heat; and the availability and nature of heat sinking and air flow. lower mosfet power calculation the calculation for heat dissipated in the lower mosfet is simple, since virtually all of the heat loss in the lower mosfet is due to current conducted through the channel resistance (r ds(on) ). in equation 14, i m is the maximum continuous output current; i pp is the peak-to-peak inductor current (see equation 1); d is the duty cycle (v out /v in ); and l is the per-channel inductance. an additional term can be added to the lower-mosfet loss equation to account for additional loss accrued during the dead time when inductor current is flowing through the lower-mosfet body diode. this term is dependent on the diode forward voltage at i m , v d(on) ; the switching frequency, f s ; and the length of dead times, t d1 and t d2 , at the beginning and the end of the lower-mosfet conduction interval respectively. thus the total maximum power dissipated in each lower mosfet is approximated by the summation of p low,1 and p low,2 . upper mosfet power calculation in addition to r ds(on) losses, a large portion of the upper- mosfet losses are due to currents conducted across the input voltage (v in ) during switching. since a substantially higher portion of the upper-mosfet losses are dependent on switching frequency, the power calculation is more complex. upper mosfet losses can be divided into separate components involving the upper-mosfet switching times; the lower-mosfet body-diode reverse- recovery charge, q rr ; and the upper mosfet r ds(on) conduction loss. when the upper mosfet turns of f, the lower mosfet does not conduct any portion of th e inductor current until the voltage at the phase node falls below ground. once the lower mosfet begins conducting, the current in the upper mosfet falls to zero as the current in the lower mosfet figure 17. voltage at iout pin with a ntc network placed between iout to ground when load current changes v_iout, 200mv/div 100a 50a 0a p low 1 , r ds on () i m n ----- - ?? ?? ?? 2 1d ? () i lpp , 2 1d ? () 12 -------------------------------- + = (eq. 14) p low 2 , v don () f s i m n ----- - i pp 2 -------- - + ?? ?? t d1 i m n ----- - i pp 2 -------- - ? ?? ?? ?? t d2 + = (eq. 15) isl6564a
22 fn6285.0 june 21, 2006 ramps up to assume the full inductor current. in equation 16, the required time for this commutation is t 1 and the approximated associated power loss is p up,1 . at turn on, the upper mosfet begins to conduct and this transition occurs over a time t 2 . in equation 17, the approximate power loss is p up,2 . a third component involves the lower mosfet?s reverse- recovery charge, q rr . since the inductor current has fully commutated to the upper mosfet before the lower- mosfet?s body diode can draw all of q rr , it is conducted through the upper mosfet across vin. the power dissipated as a result is p up,3 and is approximately finally, the resistive part of the upper mosfet?s is given in equation 19 as p up,4 . the total power dissipated by the upper mosfet at full load can now be approximated as the summation of the results from equations 16, 17, 18 and 19. since the power equations depend on mosfet parameters, choosing the correct mosfets can be an it erative process involving repetitive solutions to the loss equations for different mosfets and different switching frequencies. current sensing resistor the resistors connected between these pins and the respective phase nodes determine the gains in the load-line regulation loop and the channel-current balance loop as well as setting the overcurrent trip point. select values for these resistors based on the room temperature r ds(on) of the lower mosfets, dcr of inductor or additional resistor; the full-load operating current, i fl ; and the number of phases, n using equation 20. in certain circumstances, it ma y be necessary to adjust the value of one or more isen resistor. when the components of one or more channels are inhibited from effectively dissipating their heat so that the affected channels run hotter than desired, choose new, sma ller values of risen for the affected phases (see the section entitled channel-current balance ). choose r isen,2 in proportion to the desired decrease in temperature rise in order to cause proportionally less current to flow in the hotter phase. in equation 21, make sure that t 2 is the desired temperature rise above the ambient temperature, and t 1 is the measured temperature rise above the am bient temperature. while a single adjustment according to equation 21 is usually sufficient, it may occasionally be necessary to adjust r isen two or more times to achieve optimal thermal balance between all channels. load-line regulation resistor the load-line regulation resistor is labeled r fb in figure 8. its value depends on the desired full-load droop voltage (v droop in figure 8). if equation 20 is used to select each isen resistor, the load-line regulation resistor is as shown in equation 22. if one or more of the isen resistors is adjusted for thermal balance, as in equation 21, the load-line regulation resistor should be selected according to equation 23 where i fl is the full-load operating current and r isen(n) is the isen resistor connected to the n th isen pin. compensation the two opposing goals of compensating the voltage regulator are stability and speed. depending on whether the regulator employs the optional load-line regulation as described in load-line regulation , there are two distinct methods for achieving these goals. compensating load-line regulated converter the load-line regulated converter behaves in a similar manner to a peak-current mo de controller because the two poles at the output-filter l- c resonant frequency split with the introduction of current information into the control loop. the final location of these poles is determined by the system function, the gain of the current signal, and the value of the compensation components, r c and c c . since the system poles and zero are affected by the values of the components that are me ant to compensate them, the solution to the system equation becomes fairly complicated. fortunately there is a simple approximation that comes very close to an optimal solution. treating the system as though it were a voltage-mode regulator by compensating the l-c poles and the esr zero of t he voltage-mode approximation yields a solution that is always stable with very close to ideal transient performance. p up 1 , v in i m n ----- - i pp 2 -------- - + ?? ?? t 1 2 ---- ?? ?? ?? f s (eq. 16) p up 2 , v in i m n ----- - i pp 2 -------- - ? ?? ?? ?? t 2 2 ---- ?? ?? ?? f s (eq. 17) p up 3 , v in q rr f s = (eq. 18) p up 4 , r ds on () i m n ----- - ?? ?? ?? 2 d i pp 2 12 --------- - + (eq. 19) r isen r x 70 10 6 ? ----------------------- i fl n ------- - = (eq. 20) r isen 2 , r isen t 2 t 1 ---------- = (eq. 21) r fb v droop 70 10 6 ? ------------------------ - = (eq. 22) r fb v droop i fl r ds on () -------------------------------- r isen n () n = (eq. 23) isl6564a
23 fn6285.0 june 21, 2006 the feedback resistor, r fb , has already been chosen as outlined in load-line regulation resistor . select a target bandwidth for the co mpensated system, f 0 . the target bandwidth must be large enough to assure adequate transient performance, but smaller than 1/3 of the per-channel switching frequency. the values of the compensation components depend on the relationships of f 0 to the l-c pole frequency and the esr zero frequency. for each of the three cases which follow, there is a separate set of equations for the compensation components. in equation 24, l is the per-channel filter inductance divided by the number of active channels; c is the sum total of all output capacitors; esr is the equivalent-series resistance of the bulk output-filter capacitance; and v pp is the peak-to- peak sawtooth signal amplitude as described in figure 7 and electrical specifications . the optional capacitor c 2 , is sometimes needed to bypass noise away from the pwm comparator (see figure 18). keep a position available for c 2 , and be prepared to install a high- frequency capacitor of between 22pf and 150pf in case any leading-edge jitter problem is noted. once selected, the compensation values in equation 24 assure a stable converter with reasonable transient performance. in mo st cases, transient performance can be improved by making adjustments to r c . slowly increase the value of r c while observing the transient performance on an oscilloscope until no further improvement is noted. normally, c c will not need adjustment. keep the value of c c from equation 24 unless some performance issue is noted. compensation without load-line regulation the non load-line regulated converter is accurately modeled as a voltage-mode regulator with two poles at the l-c resonant frequency and a zero at the esr frequency. a type iii controller, as shown in figure 19, provides the necessary compensation. the first step is to choose the desired bandwidth, f 0 , of the compensated system. choose a frequency high enough to assure adequate transient per formance but not higher than 1/3 of the switching frequency. the type-iii compensator has an extra high-frequency pole, f hf . this pole can be used for added noise rejection or to assure adequate attenuation at the error-amplifier high-order pole and zero frequencies. a good general rule is to choose f hf =10f 0 , but it can be higher if desired. choosing f hf to be lower than 10f 0 can cause problems with too much phase shift below the system bandwidth. in the solutions to the compensation equations, there is a single degree of freedom. for the solutions presented in equation 24, r fb is selected arbitrarily. the remaining figure 18. compensation configuration for load-line regulated isl6564a circuit isl6564a comp c c r c r fb fb idroop vdiff - + v droop c 2 (optional) 1 2 lc ------------------- f 0 > r c r fb 2 f 0 v pp lc 0.75v in ----------------------------------- - = c c 0.75v in 2 v pp r fb f 0 ------------------------------------ = case 1: 1 2 lc ------------------- f 0 1 2 c esr () ----------------------------- - < r c r fb v pp 2 () 2 f 0 2 lc 0.75 v in -------------------------------------------- = c c 0.75v in 2 () 2 f 0 2 v pp r fb lc ------------------------------------------------------------- = case 2: (eq. 24) f 0 1 2 c esr () ----------------------------- - > r c r fb 2 f 0 v pp l 0.75 v in esr () ------------------------------------------ = c c 0.75v in esr () c 2 v pp r fb f 0 l ------------------------------------------------- = case 3: figure 19. compensation circuit for isl6564a based converter without load-line regulation isl6564a comp c c r c r fb fb idroop vdiff c 2 c 1 r 1 isl6564a
24 fn6285.0 june 21, 2006 compensation components are then selected according to equation 25. in equations 25, l is the per-channel filter inductance divided by the number of active channels; c is the sum total of all output capacitors; esr is the equivalent-series resistance of the bulk output-filter capacitance; and v pp is the peak-to-peak sawtooth signal amplitude as described in figure 7 and electrical specifications . output filter design the output inductors and the output capacitor bank together to form a low-pass filter re sponsible for smoothing the pulsating voltage at the phase nodes. the output filter also must provide the transient ene rgy until the regulator can respond. because it has a low bandwidth compared to the switching frequency, the output filter necessarily limits the system transient response. t he output capacitor must supply or sink load current while the current in the output inductors increases or decreases to meet the demand. in high-speed converters, the output capacitor bank is usually the most costly (and often the largest) part of the circuit. output filter design begins with minimizing the cost of this part of the circuit. the critical load parameters in choosing the output capacitors are the maximum size of the load step, i; the load-current slew rate, di/dt; and the maximum allowable output-volt age deviation under transient loading, v max . capacitors are characterized according to their capacitance, esr, and esl (equivalent series inductance). at the beginning of the load tr ansient, the output capacitors supply all of the transient current. the output voltage will initially deviate by an amount approximated by the voltage drop across the esl. as the load current increases, the voltage drop across the esr increases linearly until the load current reaches its final value. the capacitors selected must have sufficiently low esl and esr so that the total output- voltage deviation is less than the allowable maximum. neglecting the contribution of inductor current and regulator response, the output voltage initially deviates by an amount: the filter capacitor must have sufficiently low esl and esr so that v < v max . most capacitor solutions rely on a mixture of high-frequency capacitors with relatively lo w capacitance in combination with bulk capacitors having high capacitance but limited high-frequency performance. minimizing the esl of the high-frequency capacitors allows them to support the output voltage as the current increases. minimizing the esr of the bulk capacitors allows them to supply the increased current with less output voltage deviation. the esr of the bulk capacitors also creates the majority of the output-voltage ripple. as th e bulk capacitors sink and source the inductor ac ripple current (see interleaving and equation 2), a voltage develops across the bulk-capacitor esr equal to i c,pp (esr). thus, once the output capacitors are selected, the maximum allowable ripple voltage, v pp(max) , determines the lower limit on the inductance. since the capacitors are supplying a decreasing portion of the load current while the regulator recovers from the transient, the capacitor voltage becomes slightly depleted. the output inductors must be capable of assuming the entire load current before the output voltage decreases more than v max . this places an upper limit on inductance. equation 28 gives the upper limit on l for the cases when the trailing edge of the current transient causes a greater output-voltage deviation than the leading edge. equation 29 addresses the leading edge. normally, the trailing edge dictates the selection of l because duty cycles are usually less than 50%. nevertheless, both inequalities should be evaluated, and l should be se lected based on the lower of the two results. in each equation, l is the per-channel inductance, c is the total output capacitance, and n is the number of active channels. c c 0.75v in 2 f hf lc 1 ? ?? ?? 2 () 2 f 0 f hf lcr fb v pp ------------------------------------------------------------------ - = r c v pp 2 ?? ?? 2 f 0 f hf lcr fb 0.75 v in 2 f hf lc 1 ? ?? ?? -------------------------------------------------------------------- - = r 1 r fb c esr () lc c esr () ? ---------------------------------------- - = c 1 lc c esr () ? r fb ---------------------------------------- - = c 2 0.75v in 2 () 2 f 0 f hf lcr fb v pp ------------------------------------------------------------------ - = (eq. 25) v esl () di dt ---- -esr () i + (eq. 26) l esr () v in nv out ? ?? ?? v out f s v in v pp max () ----------------------------------------------------------- - (eq. 27) l 2ncv o i () 2 --------------------- v max i esr () ? (eq. 28) l 1.25 () nc i () 2 ------------------------- - v max iesr () ? v in v o ? ?? ?? (eq. 29) isl6564a
25 fn6285.0 june 21, 2006 input supply voltage selection the vcc input of the isl6564a can be connected either directly to a +5v supply or through a current limiting resistor to a +12v supply. an integrated 5.8v shunt regulator maintains the voltage on the vcc pin when a +12v supply is used. a 300 resistor is suggested for limiting the current into the vcc pin to a worst-case maximum of approximately 25ma. switching frequency there are a number of variables to consider when choosing the switching frequency, as there are considerable effects on the upper-mosfet loss calculation. these effects are outlined in mosfets , and they establish the upper limit for the switching frequency. the lowe r limit is established by the requirement for fast transie nt response and small output- voltage ripple as outlined in output filter design . choose the lowest switching frequency that allows the regulator to meet the transient-response requirements. switching frequency is determined by the selection of the frequency-setting resistor, r t (see the figures labeled typical application on pages 4, 5, 6 and 7). figure 20 and equation 30 are provided to assi st in selecting the correct value for r t . input capacitor selection the input capacitors are responsible for sourcing the ac component of the input current flowing into the upper mosfets. their rms current capa city must be sufficient to handle the ac component of t he current drawn by the upper mosfets which is related to duty cycle and the number of active phases. for a two phase design, use figure 21 to determine the input-capacitor rms current requirement given the duty cycle, maximum sustai ned output current (i o ), and the ratio of the per-phase peak-to-peak inductor current (i l,pp ) to i o . select a bulk capacitor with a ripple current rating which will minimize the total number of input capacitors required to support the rms current calculated. the voltage rating of the capacitors should also be at least 1.25 times greater than the maximum input voltage. figures 22 and 23 provide the same input rms current information for three and four phase designs respectively. use the same approach to selecting the bulk capacitor type and number as described above. low capacitance, high-frequency ceramic capacitors are needed in addition to the bulk capacitors to suppress leading and falling edge voltage spikes. the result from the high current slew rates produced by the upper mosfets turn on figure 20. r t vs switching frequency 100 1000 10000 10 switching frequency (khz) 10 100 1000 r t (k ) r t 10 10.886 1.0792 f s () log ? [] = (eq. 30) 0.3 0.1 0 0.2 input-capacitor current (i rms /i o ) figure 21. normalized input-capacitor rms current vs duty cycle for 2-phase converter 00.4 1.0 0.2 0.6 0.8 duty cycle (v o /v in ) i l,pp = 0 i l,pp = 0.5 i o i l,pp = 0.75 i o duty cycle (v o/ v in ) figure 22. normalized input-capacitor rms current vs duty cycle for 3-phase converter 00.4 1.0 0.2 0.6 0.8 input-capacitor current (i rms/ i o ) 0.3 0.1 0 0.2 i l,pp = 0 i l,pp = 0.25 i o i l,pp = 0.5 i o i l,pp = 0.75 i o isl6564a
26 fn6285.0 june 21, 2006 and off. select low esl ceramic capacitors and place one as close as possible to each upper mosfet drain to minimize board parasitic impedances and maximize suppression. multiphase rms improvement figure 24 is provided as a reference to demonstrate the dramatic reductions in input-c apacitor rms current upon the implementation of the multiphase topology. for example, compare the input rms current requirements of a two-phase converter versus that of a single phase. assume both converters have a duty cycle of 0.25, maximum sustained output current of 40a, and a ratio of i l,pp to i o of 0.5. the single phase converter would require 17.3 arms current capacity while the two-phase converter would only require 10.9 arms. the advantages become even more pronounced when output current is increased and additional phases are added to keep the component cost down relative to the single phase approach. layout considerations the following layout strategies are intended to minimize the impact of board parasitic impedances on converter performance and to optimize the heat-dissipating capabilities of the printed-circuit board. these sections highlight some important practices which should not be overlooked during the layout process. component placement within the allotted implementat ion area, orient the switching components first. the switching components are the most critical because they carry large amounts of energy and tend to generate high levels of noise. switching component placement should take into account power dissipation. align the output inductors and mosfets such that space between the components is minimiz ed while creating the phase plane. place the intersil mosf et driver ic as close as possible to the mosfets they control to reduce the parasitic impedances due to trace length between critical driver input and output signals. if possible, duplicate the same placement of these components for each phase. next, place the input and output capacitors. position one high-frequency ceramic input capacitor next to each upper mosfet drain. place the bulk input capacitors as close to the upper mosfet drains as dictated by the component size and dimensions. long distances between input capacitors and mosfet drains result in too much trace inductance and a reduction in ca pacitor performance. locate the output capacitors between the inductors and the load, while keeping them in close proximity to the microprocessor socket. the isl6564a can be placed off to one side or centered relative to the individual phase switching components. routing of sense lines and pwm signals will guide final placement. critical small signal components to place close to the controller include the isen resistors, r t resistor, feedback resistor, and compensation components. bypass capacitors for the isl6564a and isl66xx driver bias supplies must be placed next to their respective pins. trace parasitic impedances will reduce their effectiveness. plane allocation and routing dedicate one solid layer, usually a middle layer, for a ground plane. make all critical comp onent ground connections with vias to this plane. dedicate one additional layer for power planes; breaking the plane up into smaller islands of common voltage. use the remaining layers for signal wiring. route phase planes of copper fi lled polygons on the top and bottom once the switching component placement is set. size the trace width between the driver gate pins and the mosfet gates to carry 4a of current. when routing components in the switching path, use short wide traces to reduce the associated parasitic impedances. input-capacitor current (i rms/ i o ) figure 23. normalized input-capacitor rms current vs duty cycle for 4-phase converter 00.4 1.0 0.2 0.6 0.8 duty cycle (v o/ v in ) 0.3 0.1 0 0.2 i l,pp = 0 i l,pp = 0.25 i o i l,pp = 0.5 i o i l,pp = 0.75 i o figure 24. normalized input-capacitor rms current vs duty cycle for single-phase converter 00.4 1.0 0.2 0.6 0.8 duty cycle (v o/ v in ) input-capacitor current (i rms/ i o ) 0.6 0.2 0 0.4 i l,pp = 0 i l,pp = 0.5 i o i l,pp = 0.75 i o isl6564a
27 all intersil u.s. products are manufactured, asse mbled and tested utilizing iso9000 quality systems. intersil corporation?s quality certifications ca n be viewed at www.intersil.com/design/quality intersil products are sold by description only. intersil corpor ation reserves the right to make changes in circuit design, soft ware and/or specifications at any time without notice. accordingly, the reader is cautioned to verify that data sheets are current before placing orders. information furnishe d by intersil is believed to be accurate and reliable. however, no responsibility is assumed by intersil or its subsidiaries for its use; nor for any infringements of paten ts or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of intersil or its subsidiari es. for information regarding intersil corporation and its products, see www.intersil.com fn6285.0 june 21, 2006 isl6564a quad flat no-lead plastic package (qfn) micro lead frame pl astic package (mlfp) index d1/2 d1 d/2 d e1/2 e/2 e a 2x 0.15 b c 0.10 b a mc a n seating plane n 6 3 2 2 3 e 1 1 0.08 for odd terminal/side for even terminal/side c c section "c-c" nx b a1 c 2x c 0.15 0.15 2x b 0 ref. (nd-1)xe (ne-1)xe ref. 5 a1 4x p a c c 4x p b 2x a c 0.15 a2 a3 d2 d2 e2 e2/2 terminal tip side view top view 7 bottom view 7 5 c l c l e e e1 2 nx k nx b 8 nx l 8 8 9 area 9 4x 0.10 c / / 9 (datum b) (datum a) area index 6 area n 9 corner option 4x l1 l 10 l1 l 10 l40.6x6 40 lead quad flat no-lead plastic package (compliant to jedec mo-220vjjd-2 issue c) symbol millimeters notes min nominal max a 0.80 0.90 1.00 - a1 - - 0.05 - a2 - - 1.00 9 a3 0.20 ref 9 b 0.18 0.23 0.30 5, 8 d 6.00 bsc - d1 5.75 bsc 9 d2 3.95 4.10 4.25 7, 8 e 6.00 bsc - e1 5.75 bsc 9 e2 3.95 4.10 4.25 7, 8 e 0.50 bsc - k0.25 - - - l 0.30 0.40 0.50 8 l1 - - 0.15 10 n402 nd 10 3 ne 10 3 p- -0.609 --129 rev. 1 10/02 notes: 1. dimensioning and tolerancing conform to asme y14.5-1994. 2. n is the number of terminals. 3. nd and ne refer to the number of terminals on each d and e. 4. all dimensions are in millimeters. angles are in degrees. 5. dimension b applies to the meta llized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. the configuration of the pin #1 identifier is optional, but must be located within the zone indicated. the pin #1 identifier may be either a mold or mark feature. 7. dimensions d2 and e2 are fo r the exposed pads which provide improved electrical and thermal performance. 8. nominal dimensions are provided to assist with pcb land pattern design efforts, see intersil technical brief tb389. 9. features and dimensions a2, a3, d1, e1, p & are present when anvil singulation method is used and not present for saw singulation. 10. depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (l1) maybe present. l minus l1 to be equal to or greater than 0.3mm.

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